Phased-Array Scan Loss in mm-Wave RF Hardware

At 60° scan, projected-aperture geometry alone predicts approximately −3.01 dB. But that number is a benchmark, not a universal scan-loss limit.

Realized performance depends on the embedded element pattern. It also depends on active impedance, polarization, bandwidth, and RF electronics.

Some practical active arrays exhibit roughly 4 dB loss at 60°. [Ref. 1] . Yet carefully engineered arrays can achieve substantially lower scan loss.

A 2025 IEEE Ku-band array reported below 2 dB through ±65°. [Ref. 2]

That distinction matters.

Scan loss is not one cosine equation. It is the electromagnetic consequence of steering the complete active aperture.

First, Define What “Scan Loss” Actually Means

For realized antenna gain:

Lscan,G(θ,φ) = Grealized(0°) − Grealized(θ,φ)

For an active transmitter:

Lscan,EIRP = EIRPboresight − EIRPscan

These quantities are related, but they are not necessarily identical.

The PA output can change with scan-dependent active impedance. Feed-network loss can also vary with beam state.

Therefore, Gain scan loss ≠ automatically EIRP scan loss.

For receivers, G/T versus scan becomes equally important.

The First-Order Projected-Aperture Benchmark

Consider a flat continuous aperture. When viewed off boresight, projected area decreases approximately as:

Aprojected = A · cosθ

Therefore:

Lprojection = 10log₁₀(cosθ)

This gives:

• 30° → −0.62 dB

• 45° → −1.51 dB

• 60° → −3.01 dB

• 70° → −4.66 dB

That is an extremely useful first sanity check. But it is not the final phased-array prediction.

There Is an Important Technical Trap Here

Do not blindly add projected-aperture loss to element-pattern loss. They can represent overlapping electromagnetic physics.

For example, a cosine embedded-element model already contains angular roll-off. Adding another independent cosine term can double-count the degradation.

The safer system relationship is:

GARRAY(θ,φ) = Gembedded(θ,φ) × AF(θ,φ),

with mismatch and implementation efficiency included separately.

Analog Devices explicitly separates element factor from array factor. [Ref. 3] . That distinction becomes essential during wide-angle benchmarking.

A More Practical Scan-Loss Approximation

Qorvo models active-array scan loss approximately as:

Lscan ≈ 10log₁₀[cosⁿ(θ)],

with n typically around 1.3 for representative embedded-element behaviour. [Ref. 1]

Using n = 1.3, gives approximately:

• 30° → −0.81 dB

• 45° → −1.96 dB

• 60° → −3.91 dB

• 70° → −6.06 dB

Qorvo consequently shows approximately 2 dB at 45°. The same guide shows approximately 4 dB at 60°.

That is already materially worse than simple projected area.

What Does 4 dB Mean Systemically?

A 4 dB EIRP reduction leaves approximately 10^(−4/10) ≈ 39.8% of the boresight isotropic-equivalent power.

In a free-space link:

FSPL ∝ 20log₁₀(R)

Losing 4 dB can reduce theoretical range by approximately 37%, assuming every other link parameter remains unchanged. So wide-angle scan loss is not merely an antenna-chart detail.

It becomes a coverage-volume problem.

Key Point 1: The Element Pattern Can Dominate Wide Scan

The array factor can mathematically steer toward the horizon. The physical antenna element may not radiate efficiently there.

Analog Devices shows the element factor limiting total gain off boresight. [Ref. 3] . This is why wide-angle arrays need wide-beam embedded elements.

A brilliant beamformer cannot repair a poor element pattern.

Beamwidth Also Expands With Scan Angle

For many planar-array approximations:

HPBW(θ) ∝ 1/cosθ

Therefore, at: θ = 60°, the beamwidth approximately doubles.

Analog Devices demonstrates this foreshortening directly. [Ref. 3].

So edge-of-scan degradation has two consequences. Peak gain drops. Angular resolution also worsens.

For Radar, those are separate penalties.

Sidelobe Performance Can Degrade Simultaneously

The element pattern suppresses the scanned main beam. Some sidelobes remain closer to broadside. Their element-factor attenuation can therefore be smaller.

Analog Devices shows relative sidelobe performance degrading off boresight. [Ref. 3]

This means scan loss is not merely lost main-beam gain. It redistributes radiated energy spatially.

That matters greatly in Radar and Satellite Communications.

Polarization Can Degrade Before Gain Becomes Unacceptable

Circularly polarized arrays introduce another requirement: axial ratio versus scan angle.

A beam can retain acceptable gain. Yet polarization purity may already have deteriorated.

That directly affects polarization mismatch and cross-polar interference.

A 2026 IEEE Benchmark Shows What Good Looks Like

A recent 8×8 dual-circularly polarized array scanned multiple azimuth planes. Its active VSWR remained below 3.0 through ±60°. [Ref. 4]

Main-beam axial ratio remained below approximately 1.18 dB. Cross-polarization stayed at least 15 dB below peak copolar gain.

That is much stronger evidence than simply claiming: “Scan range = ±60°.”

Another 2026 Array Reached High Aperture Efficiency

A dual-band shared-aperture mmWave array demonstrated ±60° scanning. Its reported aperture efficiency exceeded 80% across main operating bands. [Ref. 5]

That demonstrates how element architecture can preserve wide-angle efficiency.

Wide scan is therefore fundamentally an element-plus-array co-design problem.

Key Point 2: Active Impedance Moves With Beam Angle

This is where phased arrays become active RF networks.

Each element couples electromagnetically to its neighbours. Changing excitation phase changes those coupled fields.

Therefore, the impedance seen by each PA or LNA changes.

The Active Reflection Coefficient Exposes the Mechanism

For element n:

Γactive,n = Σₘ Sₙₘ aₘ / aₙ

where:

• Sₙₘ represents mutual coupling

• aₘ represents element excitation

Change beam direction, the excitation phases change.

Therefore: Γactive changes, even if the physical PCB does not.

This is why passive single-element S₁₁ is insufficient.

Scan Angle Can Create Severe VSWR

A 2025 IEEE ISSCC study examined large mmWave phased arrays. Wide scans produced active antenna impedance variation exceeding 4:1 VSWR. [Ref. 6]

The examined conditions included scanning from 0° to 60°. Frequency also varied by approximately ±5%. A conventional PA then experienced over 7 dB power-gain variation.

That number should concern every active-array engineer.

The Passive Mismatch Penalty Is Only Part of That Problem

For: VSWR = 2:1,

the reflection coefficient magnitude is: |Γ| = 1/3

Mismatch efficiency becomes: ηm = 1 − |Γ|² ≈ 88.9%

Therefore: Lmismatch ≈ 0.51 dB

For: VSWR = 4:1,

we have: |Γ| = 0.6

and: ηm = 0.64

Therefore: Lmismatch ≈ 1.94 dB

Yet the IEEE PA study reported >7 dB gain variation. [Ref. 6].

Why?

Because the PA itself responds to the complex load trajectory.

Load Pull Becomes Scan-Angle Dependent

The active antenna load influences:

• PA gain

• OP1dB

• PAE

• AM-AM

• AM-PM

• output phase

The 2025 ISSCC work explicitly identifies these interactions. [Ref. 6]

That means EIRP scan loss can exceed antenna gain scan loss. The front-end electronics are moving with the aperture.

This Is Why Scan Blindness Occurs

At certain frequencies and scan directions, active impedance can approach severe mismatch. Array directivity then collapses.

Keysight identifies changing active impedance as a scan-blindness mechanism. [Ref. 7]

That is a much more serious condition than ordinary cosine loss.

Wide-Angle Matching Can Be Engineered

A 2024 IEEE study used artificial dielectric sheets. The technique reduced scan-dependent active reflection coefficient. [Ref. 8]

The compensated array maintained active VSWR below two.

Its scan range reached roughly:

• ±45° in E-plane

• ±65° in H-plane

• ±80° diagonally

Notice the anisotropy: A phased-array scan volume is rarely circular.

The Scan Plane Matters

“±60° scan” is incomplete information.

Was that:

• E-plane?

• H-plane?

• diagonal plane?

• every azimuth plane?

Mutual coupling differs between scan planes. The embedded element pattern also differs.

Therefore, performance boundaries are generally two-dimensional.

Key Point 3: Wide Scan Constrains Element Spacing

Scanning also changes grating-lobe conditions.

For a rectangular lattice, a useful criterion is:

d/λ < 1/[1 + sin(θmax)]

Qorvo gives this condition for grating-lobe-free scanning. [Ref. 1]

For: θmax = 60°

we obtain:

d < 0.536λ

This is much more informative than saying:

“Use half-wavelength spacing.”

At 28 GHz, That Lattice Becomes Physically Small

Free-space wavelength is approximately: λ ≈ 10.71 mm.

Therefore: dmax ≈ 0.536 × 10.71, giving: dmax ≈ 5.74 mm.

At 39 GHz, λ becomes approximately: 7.69 mm.

Therefore, dmax ≈ 4.12 mm.

Now fit inside that lattice:

• PA

• LNA

• BFIC

• bias network

• thermal path

• digital control

• RF transitions

This becomes a hardware-layout constraint.

Qorvo Identifies Exactly This mm-Wave Packaging Problem

Qorvo notes lattice spacing contracts rapidly above 28 GHz. That leaves limited space for transmit-receive electronics. [Ref. 1]

Maintaining short RF paths also preserves EIRP and receiver noise figure.

This is where antenna geometry becomes PCB architecture.

Frequency Makes Wide-Angle Scanning Harder Again

A fixed element spacing represents larger d/λ at higher frequency. Therefore, the grating-lobe margin reduces toward the upper band edge.

This is why scan analysis must use the highest operating frequency, not only centre frequency.

One layout must satisfy every frequency and scan direction.

Wideband Arrays Introduce Beam Squint

A phase shifter creates the correct delay at one frequency. The required physical delay is frequency independent.

The equivalent phase shift is not.

Therefore, wideband signals steer differently across frequency.

Measured Hardware Shows the Magnitude

Analog Devices measured a hybrid phased array calibrated at 10 GHz. A beam steered to approximately 30° shifted with frequency. [Ref. 9]

At 9 GHz, the measured peak moved near 33°. At 11 GHz, the measured peak moved near 27°.

That produces frequency-dependent scan loss at the desired direction.

True Time Delay Attacks the Correct Variable

A phase shifter controls phase.

A TTD controls propagation delay.

Analog Devices' current ADAR4001 provides true-time-delay beamforming across 2–18 GHz. ADI specifically describes its TTD architecture as beam-squint-free. [Ref. 10]

For wideband Radar, this becomes a major architecture decision.

Quantization Adds Yet Another Imperfection

Real BFICs use discrete phase states.

For B-bit phase control: ΔφLSB = 360° / 2ᴮ

For:

4 bits → 22.5°

5 bits → 11.25°

6 bits → 5.625°

Finite phase resolution produces quantization sidelobes.

Analog Devices explicitly analyzes this effect in phased arrays. [Ref. 11]

The impact grows more visible during precision beam control.

Thermal Drift Turns Scan Loss Into a Dynamic Quantity

A wide-angle beam may already have reduced margin. Now add spatial temperature gradients.

PA output changes.

BFIC phase changes.

Matching networks shift.

The aperture calibration therefore moves during operation.

This is why room-temperature boresight calibration is insufficient.

Calibration Should Be Multidimensional

For serious active arrays, characterize:

Gain = f(frequency, θ, φ, temperature, power), and

Phase = f(frequency, θ, φ, temperature, power)

The array is not one static RF network.

It contains thousands of operating states.

A Ku-Band Benchmark Shows Wide-Scan Engineering Potential

A dual-circularly polarized phased array achieved over ±60° steering. Its measured scan loss remained below 2 dB. [Ref. 2]

Axial ratio remained below 3 dB across those wide scans. A shorted conductive layer helped suppress surface waves.

This example reinforces an important point.

The −3 dB cosine benchmark is not a universal lower bound.

Element engineering can reshape the scan envelope.

A Different Ku-Band Study Used a 3 dB Performance Criterion

Another 8×8 Ku-band phased array achieved ±60° downlink scanning. The same design reached approximately ±47° uplink scanning. [Ref. 12]

Both scan loss and circular-polarization AR used a ≤3 dB criterion.

That is much more meaningful than scan angle alone.

A Recent Shared-Aperture Example Shows Another Reality

A new X/Ku shared-aperture array reported ±60° scanning. Its gain fluctuations remained within approximately 4 dB. The circular-polarization axial ratio remained below 6 dB. [Ref. 13]

This is another reminder.

Two arrays can both claim ±60° scan.

Their usable performance may differ substantially.

The Strategic Engineering Solution

Top-tier RF teams should build a scan-performance budget.

Do not treat scan angle as one antenna specification.

1. Start With the Projected-Aperture Benchmark

Calculate: 10log₁₀(cosθ).

Use it as a first-order reference.

Do not treat it as final truth.

2. Use Embedded Element Patterns

Do not use isolated-element gain alone.

Extract each element inside the active array environment.

This captures:

• edge effects

• mutual coupling

• nearby structures

• finite-array behaviour

That is the pattern the beamformer actually sees.

3. Calculate Active S-Parameters

Evaluate Γactive,n(f,θ,φ), for every important element class.

At minimum, inspect:

• central elements

• edge elements

• corner elements

They do not experience identical environments.

4. Convert Active VSWR Into PA Load Trajectories

Do not stop at S₁₁.

Take the active impedance into nonlinear PA simulation.

Evaluate:

• P1dB

• Psat

• PAE

• AM-AM

• AM-PM

• output phase

The 2025 IEEE result shows why this matters. [Ref. 6]

5. Design Lattice Spacing From Maximum Scan

Use the highest operating frequency.

For rectangular arrays:

d/λ < 1/[1 + sinθmax]

At ±60°: d < 0.536λ

This should be established before BFIC placement.

6. Design the Element for Wide Embedded Beamwidth

A narrow element pattern guarantees large scan degradation.

Options include:

• stacked patches

• magnetoelectric dipoles

• tightly coupled elements

• metasurface-assisted elements

• conductive superstrates

The correct architecture depends on bandwidth and polarization.

7. Engineer Wide-Angle Active Matching

Do not optimize only broadside S₁₁.

Optimize active VSWR versus frequency and scan.

Artificial dielectric sheets can materially improve wide-angle matching. [Ref. 8]

Other solutions include parasitic or shorted conductive structures.

8. Use True Time Delay When Bandwidth Demands It

For narrowband systems, phase shifting remains efficient.

For wideband systems, beam squint becomes increasingly expensive.

TTD aligns delay instead of one-frequency phase.

That preserves steering direction across frequency.

9. Calibrate Across Scan Angle

Do not create only one broadside calibration table.

Use Correction(f, θ, φ, T), where system complexity justifies it.

Include amplitude and phase.

For high-power arrays, include output level.

10. Include Polarization in the Scan Budget

For circularly polarized systems, measure:

• axial ratio

• copolar gain

• cross-polarization

• polarization isolation

Do not sign off from realized gain alone.

A powerful beam with poor polarization can still lose link margin.

11. Evaluate Sidelobes at Every Important Scan State

Uniform broadside sidelobes tell only one story.

Off-boresight element-factor weighting changes the pattern.

Active coupling can also change it.

For Radar, this directly influences clutter and interference susceptibility.

12. Use Full-Wave EM Before Layout Freeze

HFSS or CST Microwave Studio should evaluate:

• embedded patterns

• active impedance

• coupling

• scan blindness

• grating lobes

• surface waves

• polarization

For large arrays, combine unit-cell and finite-array modelling.

One isolated patch simulation is not array validation.

The PCB Layout Is Part of the Scan-Loss Budget

At mm-Wave, array lattice controls electronic placement.

The BFIC must fit behind the aperture.

PA routing must remain short.

Ground-via fencing must not disturb radiator currents.

Thermal structures must also fit inside the lattice.

This becomes antenna–RF–PCB–thermal co-design.

Substrate Selection Also Affects Wide Scan

Dk determines physical transmission-line geometry.

It also influences surface-wave behaviour.

Loss tangent affects feed efficiency.

Dk tolerance affects channel phase.

Copper roughness adds additional mm-Wave loss.

So laminate selection can influence the scan envelope indirectly.

Rigid-Flex RF Makes the Problem Harder

Some airborne and conformal arrays require flexible structures.

Now bending changes:

• element orientation

• mutual coupling

• polarization

• phase path

• local element spacing

A calibration measured flat may not remain valid bent.

Mechanical state therefore becomes another scan variable.

5G Infrastructure: Boresight EIRP Is Not Enough

A Massive MIMO radio serves users across sectors. Edge users rarely sit at mechanical boresight.

Therefore, procurement should request:

• EIRP versus scan

• EVM versus scan

• active return loss

• sidelobes

• thermal derating

Open RAN interoperability does not remove electromagnetic constraints.

Radar: Scan Loss Becomes Detection Loss

Radar range depends strongly on antenna gain. Wide-angle scan reduces both transmit and receive gain.

The same aperture often suffers loss in both directions.

Beam broadening simultaneously reduces angular resolution.

So edge-of-scan detection is a separate operating condition.

Satellite Communications: Low-Elevation Beams Are the Hard Ones

LEO terminals continuously steer while satellites cross the sky.

High elevation often approaches array boresight.

Low elevation demands greater electrical scan.

That is exactly where gain and polarization are most stressed.

Therefore, edge-of-scan G/T and EIRP define useful coverage.

Not broadside gain.

Aerospace & Defense: Scan Volume Must Include Environment

Temperature changes RF amplitude and phase.

Radome incidence changes with beam direction.

Structural tolerances perturb channel phase.

Platform scattering modifies embedded patterns.

MIL-STD-461 compliance addresses EMC.

It does not guarantee wide-angle array performance.

How I Would Specify a Proper Phased Array

Do not write:

“Beam scanning: ±60°.”

Specify instead:

• scan volume: azimuth and elevation

• minimum realized gain: across scan

• maximum scan loss: across frequency

• minimum EIRP: at edge scan

• minimum G/T: at edge scan

• maximum active VSWR: across scan

• maximum axial ratio: across scan

• minimum cross-polar isolation

• maximum sidelobe level

• allowed EVM: every required beam

That specification is far harder.

It is also far more meaningful.

Measurement Must Match the Specification

Modern phased-array measurement systems report much more than boresight gain.

Keysight supports EIRP versus beam scan angle. The system also reports scan range and scan loss. It can measure cross-polarization and gain compression. It also supports G/T and modulated-signal EVM measurements. [Ref. 14]

That is the correct system-level measurement philosophy.

The Summary Takeaway

The first-order projected-aperture benchmark is:

Lprojection = 10log₁₀(cosθ)

Therefore:

30° → −0.62 dB

45° → −1.51 dB

60° → −3.01 dB

70° → −4.66 dB

But practical active-array scan loss can follow stronger angular roll-off.

Qorvo's representative model gives roughly 4 dB at 60°. [Ref. 1]

At the same time, optimized IEEE arrays demonstrate below 2 dB near ±65°. [Ref. 2]

That apparent contradiction contains the real lesson:

There is no universal scan-loss number.

The actual result depends on:

• embedded element pattern

• active impedance

• mutual coupling

• lattice spacing

• polarization

• bandwidth

• PA load sensitivity

• calibration

• temperature

So the correct question is not:

“Can the array steer to ±60°?”

It is:

“What RF performance remains when the beam reaches ±60°?”

Because in a real phased array:

maximum scan angle is not the same as usable scan volume.